Drive circuit and semiconductor module utilizing a capacitance ratio between different switches

ABSTRACT

A drive circuit, including a first switch located on a low side of the drive circuit, a second switch located on a high side of the drive circuit and connected in series with the first switch, the first switch and the second switch forming an output circuit, each of the first and second switches having a high-potential end, a low-potential end and a gate terminal, a control unit that controls switching operations of the first switch and the second switch, and a capacitive element having two ends thereof respectively connected to the gate terminal of the second switch and the low potential end of the first switch.

CROSS-REFERENCE TO RELATED APPLICATION

This application is based upon and claims the benefit of priority of theprior Japanese Patent Application No. 2017-162197, filed on Aug. 25,2017, the entire contents of which are incorporated herein by reference.

BACKGROUND OF THE INVENTION 1. Field of the Invention

The embodiments discussed herein relate to a drive circuit and asemiconductor module.

2. Background of the Related Art

An output circuit including a half bridge circuit having twosemiconductor switches connected in series is generally used in a motordrive inverter and a drive circuit of a DC-DC converter of a switchingpower supply. The semiconductor switches of the output circuit are, forexample, an insulated gate bipolar transistor (IGBT) or ametal-oxide-semiconductor field-effect transistor (MOSFET).

FIG. 5 is a circuit diagram partially illustrating an LLC currentresonant DC-DC converter.

This DC-DC converter includes a drive circuit including a main controlcircuit 102 and main switches Q101 and Q102 connected in series to adirect-current power supply 101, as well as an inductor L101, atransformer T101, a capacitor C101, and a direct-current voltage outputcircuit 103. Here, the main switches Q101 and Q102 are MOSFETs and thusinclude body diodes D101 and D102 connected in antiparallel to theseMOSFETs, respectively.

The main control circuit 102 is connected to the gate terminal of thelow-side main switch Q101 and the gate terminal of the high-side mainswitch Q102. The main control circuit 102 controls the main switchesQ101 and Q102 to turn on and off alternatingly. A resonance circuitincluding the inductor L101, the primary winding of the transformerT101, and the capacitor C101 connected in series is connected to acommon connection point between the main switches Q101 and Q102. Thedirect-current voltage output circuit 103 for rectification andsmoothing is connected to the secondary winding of the transformer T101,and outputs and supplies a direct-current voltage to a load.

In the above configuration, when the low-side main switch Q101 is turnedoff and the high-side main switch Q102 is turned on, the electriccurrent supplied from the positive terminal of the direct-current powersupply 101 returns to the negative terminal of the direct-current powersupply 101 through the main switch Q102 and the resonance circuit. Then,when both of the main switches Q101 and Q102 are turned off, theelectric current of the resonance circuit flows through the body diodeD101 of the main switch Q101 in order to continue the electric currentin the inductor L101 and the primary winding of the transformer T101. Inthis case, the change (dV/dt) of the drain-source voltage and the change(dI/dt) of the drain current are generated in the main switch Q101,without ON operation of the main switch Q101.

Thereafter, when the low-side main switch Q101 is turned on before theresonant current flowing through the resonance circuit reverses itsdirection, the electric current of the resonance circuit flows throughthe main switch Q101. Then, when both of the main switches Q101 and Q102are turned off, the resonant current flows through the body diode D102of the main switch Q102 to continue the electric current in the inductorL101 and the primary winding of the transformer T101, and therebyre-generates and accumulates electric power in the direct-current powersupply 101.

By repeating the above operation of the main switches Q101 and Q102,alternate-current voltage generated in the secondary winding of thetransformer T101 is rectified and smoothed in the direct-current voltageoutput circuit 103, which outputs direct-current voltage of apredetermined magnitude.

Here, after electric current changes its direction from the negativedirection (commutation current state) to the positive direction in oneof the main switches Q101 and Q102, the one of the main switches isturned off to flow the electric current to the other of the mainswitches Q101 and Q102, and then the other main switch is turned onwhile the electric current flows in the negative direction in the othermain switch. In this manner, a large change of the electric potential isnot applied to the body diodes D101 and D102 at the time of turning on.However, when the load of the converter or the input voltage rapidlychanges (for example, from an overload state to an unloaded state), whenthe power supply starts operating, or in like cases, the main controlcircuit 102 of the converter is unable to follow such a change and turnson the main switch Q102 while the electric current flows through themain switch Q101 in the negative direction in the commutation currentstate. In this case, the voltage of the direct-current power supply 101is instantaneously applied to the body diode D101 of the main switchQ101 in a reverse bias state, and the body diode D101 performs recoveryoperation. In this recovery operation, rapid change dI/dt of theelectric current (referred to here as “resonance loss”) occurs, possiblydestroying the body diode D101.

In the past, the main control circuit 102 has been improved to preventthe rapid change dI/dt of the electric current (for example, refer toJapanese Patent No. 5761206). In Japanese Patent No. 5761206, the maincontrol circuit has a resonance loss prevention function. This resonanceloss prevention function monitors the currents or voltages of the mainswitches and turns on the main switch that is not the main switchthrough which electric current flows in the negative direction, at atime point that does not cause a resonance loss.

Moreover, there is a method in which gate drive voltage is generated ina direction canceling the rapid current change dI/dt that occurs in amain circuit, by magnetically coupling a gate drive terminal and aterminal through which the main circuit current flows (for example,refer to Japanese Patent No. 6065744).

Furthermore, there is a method that increases the gate resistance inproportion to the main circuit current to reduce the effective forwardtransfer conductance (gfs) of a main switch, and thereby reduces theperformance of the drive circuit of the other switch (for example, referto Japanese Laid-open Patent Publication No. 2013-110878). That is, inJapanese Laid-open Patent Publication No. 2013-110878, even if electriccurrent continues flowing in the negative direction in one of theswitches when the other switch is turned on, the forward transferconductance of the other switch is low enough to reduce the change dI/dtof the electric current during recovery operation.

However, the technologies described in Japanese Patent No. 5761206 andJapanese Laid-open Patent Publication No. 2013-110878 have a problem ofcomplexity of the drive circuit, because of additional circuits, such asa function for interpreting the state of the main electric current.Moreover, the technology described in Japanese Patent No. 6065744 has aproblem of difficulty in circuit layout, because the control terminal(gate drive terminal) is wired along the main circuit to couple theterminals magnetically.

SUMMARY OF THE INVENTION

According to one aspect, there is provided a drive circuit including: afirst switch located on a low side; a second switch located on a highside and connected in series with the first switch, the first switch andthe second switch forming an output circuit; a main control unit thatcontrols switching operation of the first switch and the second switch;and a first capacitive element having two ends connected between a gateterminal of the second switch and a low potential end of the firstswitch.

The object and advantages of the invention will be realized and attainedby means of the elements and combinations particularly pointed out inthe claims.

It is to be understood that both the foregoing general description andthe following detailed description are exemplary and explanatory and arenot restrictive of the invention.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram illustrating a configuration example of aDC-DC converter that employs a drive circuit according to a firstembodiment.

FIGS. 2A and 2B are diagrams for describing operation of a drive circuitaccording to the first embodiment; FIG. 2A illustrates an output circuitof a drive circuit; and FIG. 2B illustrates behavior performed when aresonance loss occurs.

FIG. 3 is a circuit diagram illustrating a configuration example of aDC-DC converter that employs a drive circuit according to a secondembodiment.

FIGS. 4A and 4B are diagrams for describing operation of a drive circuitaccording to the second embodiment; FIG. 4A illustrates an outputcircuit of a drive circuit; and FIG. 4B illustrates behavior performedwhen a resonance loss occurs.

FIG. 5 is a circuit diagram partially illustrating an LLC currentresonant DC-DC converter.

DETAILED DESCRIPTION OF THE INVENTION

Several embodiments will be described below with reference to theaccompanying drawings, wherein like reference numerals refer to likeelements throughout. These embodiments describe application examples ofan LLC current resonant DC-DC converter. Note that each embodiment maybe carried out by partially combining a plurality of embodiments withoutinconsistency.

First Embodiment

FIG. 1 is a circuit diagram illustrating a configuration example of aDC-DC converter that employs a drive circuit according to a firstembodiment. FIGS. 2A and 2B are diagrams for describing operation of thedrive circuit according to the first embodiment. FIG. 2A illustrates anoutput circuit of the drive circuit. FIG. 2B illustrates the behaviorperformed when a resonance loss occurs.

A DC-DC converter includes a low-side switching element Q1 and ahigh-side switching element Q2 as main switches. The switching elementsQ1 and Q2 are connected in series to form a half-bridge output circuit.In the illustrated example, the switching elements Q1 and Q2 areN-channel MOSFETs and thus include body diodes D1 and D2 connected inantiparallel, respectively. The high potential ends of the switchingelements Q1 and Q2 are drain terminals, and the low potential ends ofthe switching elements Q1 and Q2 are source terminals. Note that themain switches are not limited to N-channel MOSFETs but may employ IGBTs.In the case of IGBTs, the main switches are each configured such that afreewheeling diode (FWD) is connected in antiparallel to an IGBT, orsuch that an IGBT and an FWD are integrated to form a reverse conductingIGBT (RC-IGBT).

The gate terminals of the switching elements Q1 and Q2 and a commonconnection point between the switching elements Q1 and Q2 are connectedto a main control circuit 10. The main control circuit 10 includes alow-side control circuit 11, a low-side drive circuit 12, a high-sidecontrol circuit 13, and a high-side drive circuit 14. The output of thelow-side control circuit 11 is connected to the input of the low-sidedrive circuit 12, and the output of the low-side drive circuit 12 isconnected to the gate terminal of the switching element Q1. The outputof the high-side control circuit 13 is connected to the input of thehigh-side drive circuit 14, and the output of the high-side drivecircuit 14 is connected to the gate terminal of the switching elementQ2.

The gate terminal of the high-side switching element Q2 is connected toa terminal of a capacitor (first capacitive element) C1, and the otherterminal of the capacitor C1 is connected to the source terminal (lowpotential end) of the low-side switching element Q1.

The drain terminal of the high-side switching element Q2 is connected tothe positive terminal of a direct-current power supply 20, and thesource terminal of the low-side switching element Q1 is connected to thenegative terminal of the direct-current power supply 20. Thedirect-current power supply 20 may be a power factor corrector circuitthat converts and boosts the alternate-current voltage of a commercialpower supply to direct-current voltage, for example.

The common connection point between the switching elements Q1 and Q2 isconnected to a terminal of an inductor 31, and the other terminal of theinductor 31 is connected to a terminal of a primary winding 33 of atransformer 32. The other terminal of the primary winding 33 isconnected to a terminal of a resonance capacitor 34, and the otherterminal of the resonance capacitor 34 is connected to the sourceterminal of the low-side switching element Q1. Here, a resonance circuitis formed of the inductor 31, the resonance capacitor 34, and a leakageinductance component between the primary winding 33 and secondarywindings 35 and 37 of the transformer 32.

A terminal of the secondary winding 35 of the transformer 32 isconnected to the anode terminal of a diode 36, and a terminal of thesecondary winding 37 is connected to the anode terminal of a diode 38.The cathode terminals of the diodes 36 and 38 are connected together toan output terminal 40 p and the positive terminal of an output capacitor39. The negative terminal of the output capacitor 39 is connected to anoutput terminal 40 n and a common connection point between the secondarywindings 35 and 37. The secondary windings 35 and 37, the diodes 36 and38, and the output capacitor 39 form a direct-current voltage outputcircuit of the DC-DC converter that rectifies and smooths thealternate-current voltage generated in the secondary windings 35 and 37to convert the alternate-current voltage to direct-current voltage andoutputs the direct-current voltage.

The positive terminal of the output capacitor 39 is connected to themain control circuit 10 via a feedback circuit 41. The feedback circuit41 detects the voltage between the terminals of the output capacitor 39,and feeds information of an error between the detected voltage and atarget output voltage, back to the main control circuit 10. The maincontrol circuit 10 controls the switching elements Q1 and Q2 to adjustthe voltage between the terminals of the output capacitor 39 to thetarget output voltage on the basis of the error information.

Here, the main control circuit 10 is configured with a monolithicintegrated circuit. The drive circuit including the main control circuit10, the switching elements Q1 and Q2, and the capacitor C1 is containedin one package, which is a semiconductor module 50 called intelligentpower module (IPM). In the semiconductor module 50, the capacitor C1 isprovided outside the main control circuit 10 as illustrated in FIG. 1,but may be formed in the monolithic integrated circuit of the maincontrol circuit 10.

Next, the operation of the DC-DC converter having the aboveconfiguration will be described. First, the main control circuit 10alternatingly turns on and off the switching elements Q1 and Q2, bymeans of the low-side control circuit 11, the low-side drive circuit 12,the high-side control circuit 13, and the high-side drive circuit 14.

That is, the LLC current resonant DC-DC converter has first to fourthoperation states described below, and controls resonant current byrepeating these four operation states.

In the first operation state, the low-side switching element Q1 isturned off, and the high-side switching element Q2 is turned on. In thisfirst operation state, resonant current flows through the switchingelement Q2 and the resonance circuit in the positive direction,energized by the direct-current power supply 20.

In the second operation state, the switching element Q2 is turned offwhile electric current flows through the switching element Q2 in thepositive direction. In this second operation state, the switchingelement Q2 is turned off, and immediately thereafter the electriccurrent becomes a commutation current state in which the electriccurrent flows through the body diode D1 of the switching element Q1 inthe negative direction toward the switching element Q2, and the resonantcurrent continuously changes in the positive direction. The low-sideswitching element Q1 is turned on at a time point when the voltage(drain-source voltage) of the switching element Q1 becomes approximatelyzero while the electric current flows through the body diode D1.

In the third operation state, the electric current flows through theswitching element Q1 in the positive direction, when the resonantcurrent changes its direction from the positive direction to thenegative direction while the low-side switching element Q1 is turned on.

In the fourth operation state, the switching element Q1 is turned off,while the electric current flows through the switching element Q1 in thepositive direction. In this fourth operation state, the switchingelement Q1 is turned off, and immediately thereafter the electriccurrent flows through the body diode D2 of the switching element Q2 inthe negative direction toward the direct-current power supply 20, andthe resonant current continuously changes in the negative direction.Also, the high-side switching element Q2 is turned on at a time pointwhen the voltage (drain-source voltage) of the switching element Q2becomes approximately zero while the electric current flows through thebody diode D2.

The switching elements Q1 and Q2 are turned on and off alternatingly,and thereby the resonance circuit (including the inductor 31, theleakage inductance component of the transformer 32, and the resonancecapacitor 34) performs resonance behavior, so that resonant current isinduced in the secondary windings 35 and 37 of the transformer 32. Thisinduced current is rectified and smoothed by the diodes 36 and 38 andthe output capacitor 39 and is output from the output terminals 40 p and40 n as direct-current output voltage.

The output voltage output from the output terminals 40 p and 40 n isdetected by the feedback circuit 41, and the feedback circuit 41 feedsthe information of an error between the detected voltage and the targetoutput voltage, back to the main control circuit 10. Upon receiving theerror information, the main control circuit 10 controls and modulatesthe pulse widths applied to the switching elements Q1 and Q2 in such amanner to adjust the output voltage to the target output voltage.

Next, an effect of the capacitor C1 provided between the gate terminalof the high-side switching element Q2 and the low potential end of thelow-side switching element Q1 will be described. In the output circuitof FIG. 2A, a gate capacitance Cgs(Q2), which is an input capacitance,is illustrated in the high-side switching element Q2 for the purpose ofdescribing the operation. FIG. 2B illustrates the gate voltage Vgs(Q2),the drain current I(Q2), and the drain-source voltage Vds(Q2) of thehigh-side switching element Q2, and the drain-source voltage Vds(Q1) andthe drain current I(Q1) of the low-side switching element Q1, from thetop of FIG. 2B. Note that the positive direction of the drain currentI(Q1) of the low-side switching element Q1 is the direction flowing fromthe source terminal to the drain terminal.

First, in the second operation state of the LLC current resonant DC-DCconverter, the electric current I(Q1) flows through the body diode D1 ofthe switching element Q1 in the negative direction, immediately afterthe switching element Q2 is turned off. In this state, the switchingelement Q2 starts ON operation, when the operation of the main controlcircuit 10 is unable to follow the rapid change of the load and theinput voltage and the gate voltage Vgs(Q2) exceeds a threshold value toturn on the switching element Q2. Upon the start of ON operation, thedrain-source voltage Vds(Q1) of the switching element Q1 is rapidlysubject to reverse bias to generate sharp turn-off dv/dt. A part ofelectric charge generated by this dv/dt forms a transient electriccurrent that flows a loop starting from the drain of the switchingelement Q1 via the gate capacitance Cgs(Q2) of the switching element Q2and the capacitor C1 and returning to the source of the switchingelement Q1. This electric current flows through the gate capacitanceCgs(Q2) of the switching element Q2 and the capacitor C1, and therebythe gate voltage Vgs(Q2) applied to the gate terminal of the switchingelement Q2 by the main control circuit 10 is reduced by the drain-sourcevoltage Vds(Q1) of the switching element Q1 divided by using the gatecapacitance Cgs(Q2) of the switching element Q2 and the capacitor C1. Asa result, the change curve of the gate voltage Vgs(Q2) of the switchingelement Q2 is reduced from the curve illustrated with a solid line tothe curve illustrated with a dashed line, as illustrated in FIG. 2B. Asdescribed above, the gate voltage Vgs(Q2) of the switching element Q2having started turn-on behavior is reduced, and thus the characteristicsof the drain current I(Q2) of the switching element Q2 is also reducedfrom the solid line to the dashed line as illustrated in FIG. 2B.Thereby, at the body diode D1 of the switching element Q1, the change(dI/dt) of the electric current I(Q1) is reduced during the recoverytime.

Note that the drain-source voltage Vds(Q1) of the switching element Q1is applied to the gate capacitance Cgs(Q2) of the switching element Q2and the capacitor C1, and the voltage obtained by dividing thedrain-source voltage Vds(Q1) is applied to the gate capacitance Cgs(Q2)of the switching element Q2. Hence, consideration is to be given toprevention of a situation in which the gate voltage Vgs(Q2) of theswitching element Q2 exceeds a breakdown voltage between the gate andthe source and destroys the switching element Q2.

That is, the gate voltage Vgs(Q2), which is the gate-source voltage ofthe switching element Q2, is expressed by the drain-source voltageVds(Q1) of the switching element Q1 and a capacitance division ratiobetween the gate capacitance Cgs(Q2) of the switching element Q2 and thecapacitor C1.−Vgs(Q2)=Vds(Q1)×C1/(C1+Cgs(Q2))  (1)

In the following, Vgs(Q2)bd represents the breakdown voltage (allowablevoltage) between the gate and the source of the switching element Q2,and Vds(Q1)max represents the maximum possible voltage applied betweenthe drain and the source of the switching element Q1. In order toprevent the switching element Q2 from exceeding the breakdown voltagebetween the gate and the source,|Vgs(Q2)|<Vgs(Q2)bd  (2)is to be established. When equation (1) is incorporated into thisinequality (2),Vgs(Q2)bd>Vds(Q1)×C1/(C1+Cgs(Q2))  (3)is established, and this inequality (3) is transformed toC1/(C1+Cgs(Q2))<Vgs(Q2)bd/Vds(Q1)max  (4).That is, the ratio of the capacitance (first capacitance) of thecapacitor C1 to the capacitance (fourth capacitance) obtained by addingthe gate capacitance Cgs(Q2) (third capacitance) of the switchingelement Q2 to the capacitance of the capacitor C1 is preferably smallerthan the ratio of the gate breakdown voltage of the switching element Q2to the maximum possible voltage applied between the drain and the sourceof the switching element Q1.

Second Embodiment

FIG. 3 is a circuit diagram illustrating a configuration example of aDC-DC converter that employs a drive circuit according to a secondembodiment. FIGS. 4A and 4B are diagrams for describing the operation ofthe drive circuit according to the second embodiment. FIG. 4Aillustrates an output circuit of the drive circuit. FIG. 4B illustratesthe behavior performed when a resonance loss occurs. In FIG. 3, thecomponents that are the same as or equivalent to the componentsillustrated in FIG. 1 are denoted with the same reference signs, andtheir detailed description will be omitted.

In the second embodiment, a terminal of a capacitor (second capacitiveelement) C2 is connected to the gate terminal of the low-side switchingelement Q1, and the other terminal of the capacitor C2 is connected tothe source terminal, which is the low potential end, of the high-sideswitching element Q2. Components except this capacitor C2 are the sameas those in the first embodiment. Thus, in the following, the behaviorrelevant to the added capacitor C2 is described mainly. Note that thiscapacitor C2 is provided outside the main control circuit 10 asillustrated in FIG. 3, but may be formed in a monolithic integratedcircuit being the main control circuit 10.

Here, the effect of the capacitor C2 will be described. In the outputcircuit illustrated in FIG. 4A, the gate capacitance Cgs(Q1) of thelow-side switching element Q1 is illustrated. FIG. 4B illustrates thedrain-source voltage Vds(Q2) and the drain current I(Q2) of thehigh-side switching element Q2, and the drain current I(Q1), thedrain-source voltage Vds(Q1), and the gate voltage Vgs(Q1) of thelow-side switching element Q1, from the top of FIG. 4B. Note that thepositive direction of the drain current I(Q2) of the high-side switchingelement Q2 is the direction flowing from the source terminal to thedrain terminal.

Here, the electric current I(Q2) flows through the body diode D2 of thehigh-side switching element Q2 in the negative direction, immediatelyafter the switching element Q1 is turned off, as in the fourth operationstate of the LLC current resonant DC-DC converter. In this state, theswitching element Q1 is turned on, when the main control circuit 10 isunable to follow the rapid change of the load and the input voltage andthe gate voltage Vgs(Q1) exceeds a threshold value to turn on theswitching element Q1. Thereby, the conductive body diode D2 isinstantaneously subject to reverse bias, and the drain-source voltageVds(Q2) of the switching element Q2 is rapidly subject to reverse biasto generate rapid turn-off dv/dt, and simultaneously turn-on dv/dt isgenerated in the drain-source voltage Vds(Q1) of the switching elementQ1. This dv/dt causes the electric charge to move from the source of theswitching element Q1 via the gate capacitance Cgs(Q1) of the switchingelement Q1 and the capacitor C2 to the drain of the switching elementQ1, so as to flow the electric current in this loop. As a result, thegate voltage Vgs(Q1) maintained by the gate capacitance Cgs(Q1) of theswitching element Q1 is lowered transiently, and the change (+dI/dt) ofthe electric current of the switching element Q2 is reducedconsequently.

Moreover, the capacitor C2 connected to the gate terminal of thelow-side switching element Q1 needs to be set to a capacitance valuethat does not destroy the switching element Q1. The gate voltageVgs(Q1), which is the voltage between the gate and the source of theswitching element Q1, is expressed by the drain-source voltage Vds(Q1)of the switching element Q1 and a capacitance division ratio between thegate capacitance Cgs(Q1) of the switching element Q1 and the capacitorC2.Vgs(Q1)=Vds(Q1)×C2/(C2+Cgs(Q1))  (5)

In the following, Vgs(Q1)bd represents the breakdown voltage (allowablevoltage) between the gate and the source of the switching element Q1,and Vds(Q1)max represents the maximum possible voltage applied betweenthe drain and the source of the switching element Q1. In this case, inorder to prevent the switching element Q1 from exceeding the breakdownvoltage between the gate and the source,|Vgs(Q1)|<Vgs(Q1)bd  (6)is to be established. When equation (5) is incorporated into thisinequality (6),Vgs(Q1)bd>Vds(Q1)×C2/(C2+Cgs(Q1))  (7)is established, and this inequality (7) is transformed toC2/(C2+Cgs(Q1))<Vgs(Q1)bd/Vds(Q1)max  (8).That is, the ratio of the capacitance (second capacitance) of thecapacitor C2 to the capacitance (sixth capacitance) obtained by addingthe gate capacitance Cgs(Q1) (fifth capacitance) of the switchingelement Q1 to the second capacitance is preferably smaller than theratio of the gate breakdown voltage of the switching element Q1 to themaximum possible voltage applied between the drain and the source of theswitching element Q1.

Moreover, the capacitance value of the capacitor C2 is preferably equalto the capacitance value of the capacitor C1. Thereby, the property ofthe high side of the output circuit is identical with the property ofthe low side of the output circuit.

In the above, the preferable embodiments have been described, but thepresent disclosure is not limited to the above specific embodiments. Forexample, this drive circuit may be applied to a drive circuit fordriving an inductive load, such as a motor.

The drive circuit and the semiconductor module of the aboveconfiguration have an advantage that the first capacitive elementreduces the gate voltage supplied to the gate terminal of the secondswitch when the electric current flows through the first switch in thenegative direction, and thus even if the second switch is turned on, itsresultant change is reduced so as not to destroy the first switch.

All examples and conditional language provided herein are intended forthe pedagogical purposes of aiding the reader in understanding theinvention and the concepts contributed by the inventor to further theart, and are not to be construed as limitations to such specificallyrecited examples and conditions, nor does the organization of suchexamples in the specification relate to a showing of the superiority andinferiority of the invention. Although one or more embodiments of thepresent invention have been described in detail, it should be understoodthat various changes, substitutions, and alterations could be madehereto without departing from the spirit and scope of the invention.

What is claimed is:
 1. A drive circuit comprising: a first switchlocated on a low side of the drive circuit; a second switch located on ahigh side of the drive circuit and connected in series with the firstswitch, the first switch and the second switch forming an outputcircuit, each of the first and second switches having a high-potentialend, a low-potential end and a gate terminal; a control unit thatcontrols switching operations of the first switch and the second switch;and a capacitive element having two ends thereof respectively connectedto the gate terminal of the second switch and the low potential end ofthe first switch, the capacitive element being of a first capacitance,wherein a ratio of the first capacitance to a sum of the firstcapacitance and a third capacitance, the third capacitance being acapacitance between the gate terminal of the second switch and the lowpotential end of the second switch, is smaller than a ratio of a gatebreakdown voltage of the second switch to a voltage applied between thehigh potential end of the first switch and the low potential end of thefirst switch.
 2. The drive circuit according to claim 1, furthercomprising: another capacitive element having two ends thereofrespectively connected to the gate terminal of the first switch and thelow potential end of the second switch, said another capacitive elementbeing of a second capacitance.
 3. The drive circuit according to claim2, wherein the first capacitance is equal to the second capacitance. 4.The drive circuit according to claim 2, wherein a ratio of the secondcapacitance to a sum of the second capacitance and a fifth capacitance,the fifth capacitance being a capacitance between the gate terminal ofthe first switch and the low potential end of the first switch, issmaller than a ratio of a gate breakdown voltage of the first switch tothe voltage applied between the high potential end of the first switchand the low potential end of the first switch.
 5. The semiconductormodule according to claim 1, wherein the control unit is a monolithicintegrated circuit, and at least one of the first capacitive element andthe second capacitive element is formed in the monolithic integratedcircuit.
 6. The semiconductor module according to claim 1, wherein thecontrol unit is a monolithic integrated circuit, and at least one of thefirst capacitive element and the second capacitive element is providedin a module.
 7. A semiconductor module comprising: a first switchlocated on a low side of the semiconductor module; a second switchlocated on a high side of the semiconductor module and connected inseries with the first switch, the first switch and the second switchforming an output circuit, each of the first and second switches havinga high-potential end, a low-potential end and a gate terminal; a controlunit that controls switching operations of the first switch and thesecond switch; a first capacitive element having two ends thereofrespectively connected to the gate terminal of the second switch and thelow potential end of the first switch, the first capacitive elementbeing of a first capacitance, wherein a ratio of the first capacitanceto a sum of the first capacitance and a third capacitance, the thirdcapacitance being a capacitance between the gate terminal of the secondswitch and the low potential end of the second switch, is smaller than aratio of a gate breakdown voltage of the second switch to a voltageapplied between the high potential end of the first switch and the lowpotential end of the first switch; and a second capacitive elementhaving two ends thereof respectively connected to the gate terminal ofthe first switch and the low potential end of the second switch, thesecond capacitive element being of a second capacitance, wherein a ratioof the second capacitance to a sum of the second capacitance and a fifthcapacitance, the fifth capacitance being a capacitance between the gateterminal of the first switch and the low potential end of the firstswitch, is smaller than a ratio of a gate breakdown voltage of the firstswitch to the voltage applied between the high potential end of thefirst switch and the low potential end of the first switch, and thefirst capacitance of the first capacitive element is equal to the secondcapacitance of the second capacitive element.
 8. The semiconductormodule according to claim 7, wherein the control unit is a monolithicintegrated circuit, and at least one of the first capacitive element andthe second capacitive element is formed in the monolithic integratedcircuit.
 9. The semiconductor module according to claim 7, wherein thecontrol unit is a monolithic integrated circuit, and at least one of thefirst capacitive element and the second capacitive element is providedin a module.